Portable wireless communication devices, such as cellular handsets, personal digital assistants (PDAs) and hand-held computers, are becoming smaller and lighter with each new generation of wireless communication technology. They are also becoming more technically sophisticated, and currently often provide or support, in addition to traditional voice communications, features such as built-in cameras, Bluetooth connectivity, text and instant messaging, and mini browsers for surfing the Internet. These additional features can severely limit the device's battery life.
To address this problem, major efforts have been directed at ways to improve the power efficiencies of these types of devices. Some of these efforts have included researching and discovering new rechargeable battery chemistries that offer longer battery life cycles. Others have focused on improving the efficiencies of the electrical circuits which use the battery's power. Because the radio frequency (RF) power amplifier (PA) output circuitry (i.e., the RF PA) in portable wireless communication devices is often the circuit that consumes the most battery power, efforts to improve circuit efficiencies have largely focused on improving the efficiency of the RF PA circuitry. Unfortunately, because conventional power amplifier circuits must operate linearly, but are not very efficient when configured to do so, improving the power efficiency of conventional RF PAs has been a very difficult problem.
Modern wireless communication standards, such as EDGE (Enhanced Data rates for GSM (Global System for Mobile Communications) Evolution) and W-CDMA (Wideband Code Division Multiple Access) employ non-constant envelope signals. To minimize distortion of these types of signals (e.g., to prevent signal peak clipping), the RF PA must be configured for linear operation. This requires the drive levels to the RF PA to be reduced, and, depending on the crest factor level of the signal (i.e., the peak amplitude of the signal divided by the root mean square (RMS) value of the signal), additional linearization resources may be required to ensure signal integrity. The immediate consequence of this linearization effort is a reduction in efficiency.
Another type of transmitter, known as a polar modulation transmitter, avoids the linearity-efficiency tradeoff of conventional power amplifiers. Because of its superior efficiency characteristics, its adaptability to different modulation schemes, and its ability to process state-of-the-art non-constant envelope communications signals, such as EDGE and W-CDMA, the polar modulation transmitter has gained widespread use in recent years.
FIG. 1 is a block diagram of a typical polar modulation transmitter 100. The polar modulation transmitter 100 comprises a symbol generator 102; a rectangular-to-polar converter 104; an envelope path including an envelope digital to analog converter (DAC) 106 and an envelope modulator circuit 108; a phase path including a phase path DAC 110 and a VCO 112; an RF PA 114; and an antenna 116.
The polar modulation transmitter 100 operates by first receiving a digital message at the symbol generator 102. Using the digital data in the digital message, the symbol generator 102 generates in-phase (‘I’ phase) and quadrature phase (‘Q’ phase) baseband signals. These I and Q baseband signals are received by the rectangular-to-polar converter 104, which, as the name suggests, converts the I and Q baseband signals into amplitude (i.e., ‘envelope’) and phase component signals, as indicated by the ‘ρ’ and ‘θ’ symbols in FIG. 1, respectively.
The phase path DAC 110 operates to convert the phase component signal into an analog waveform, which drives the VCO 112 to create a phase modulated RF carrier signal (i.e., ‘PM’ signal). Meanwhile, the envelope DAC 106 operates to convert the envelope component signal (i.e., the amplitude modulation or ‘AM’ signal) into an analog waveform. This analog envelope component signal is coupled to the envelope modulator circuit 108, which operates to modulate a power supply voltage, VBATT (e.g., as provided by the wireless communication device's battery corresponding to a DC supply 118), according to variations in amplitude of the envelope signal. In this manner an amplitude modulated power supply signal, VS, is created.
To generate the final modulated RF carrier signal which the antenna 116 can radiate over the air, the amplitude modulated power supply signal from the envelope path is coupled to the power supply port of the RF PA 114 while the RF PM signal is coupled to the RF input port of the RF PA 114. The RF PA 114 operates to superimpose the envelope information onto the RF signal at the output port of the RF PA 114. Because the peak amplitude of the signal into the RF PA 114 remains constant over time, the linearity concerns involving amplifying non-constant envelope signals are avoided. For this reason more efficient, non-linear RF PAs such as, for example, Class-D and E switch-mode RF PAs, can be used.
FIG. 2 is a simplified drawing illustrating how the output stage of a polar modulation transmitter may be implemented using a switch-mode RF PA. A switch-mode RF PA is formed using one or more active devices (e.g., bipolar junction or field-effect transistors). The RF phase modulated signal (‘RF IN’ in the drawing) is used to control the opening and closing of a switch-mode RF PA 114. As described above, the envelope portion of the baseband signal (i.e., the ‘AM’ signal) is used by the envelope modulator circuit 108 to amplitude modulate the DC supply 118, which usually comprises the rechargeable battery of an associated wireless communications device.
Typically, the battery voltage, VBATT, provided by the DC supply 118 is about 3 to 3.5 volts (up to around 4.2 volts when fully charged). Yet cellular networks, such as the widely deployed GSM cellular network, require output powers of 3 Watts or higher. Given this large output power, but limited supply voltage, and that the power consumed by the RF PA is proportional to the square of the amplitude of the amplitude modulated power supply voltage, VS, applied to it, it is important that only a small portion of the battery voltage, VBATT, be allowed to drop across the envelope modulator circuit 108.
FIG. 3 is a drawing of a prior art envelope modulator circuit 108 which is commonly used to maximize the transfer of the modulated battery voltage to the RF PA (represented by a load resistance 114 having a resistance, RPA, in the drawing). The envelope modulator circuit 108 comprises an operational amplifier (op-amp) 302 configured as an inverting amplifier that drives the base of a p-n-p bipolar junction transistor (BJT) 304. During times when the power required of the RF PA 114 is high, the p-n-p BJT 304 is configured so that it operates near saturation. When operating near saturation the collector-emitter voltage VCE(sat) of the p-n-p BJT 304 is only on the order of about 0.1 volts. Consequently, most of the available battery supply voltage, VBATT, is made available at the power supply port of the RF PA 114 and not dropped across the envelope modulator circuit 108. In other words, VS=VBATT−VCE(sat)≅VBATT.
While the envelope modulator circuit 108 in FIG. 3 does work efficiently when the RF PA 114 must provide high output powers, it does have various drawbacks when only low output power levels are needed. These drawbacks relate in particular to use of the p-n-p type BJT 304, which when operating at low powers can be susceptible to noise. This susceptibility to noise is highly undesirable, since noise can cause distortions in the RF output signal of the RF PA 114.
The p-n-p BJT 304, particularly when combined with the inverting op-amp of the envelope modulator circuit of FIG. 3, also presents stability concerns. Variation in the amplitude of the AM signal applied to the input of the op-amp 302 can cause wide variations in the transistor collector current, IC. This wide variation in collector current demands a corresponding wide variation in closed loop gain of the amplifier. However, the gain of practical op-amps is frequency dependent, and decreases and approaches unity as the frequency of its input signal (in this case, the AM signal) increases. The phase shift between the AM signal and the signal fed back to the noninverting input of the op-amp 302 also increases towards 180° as the frequency of the AM signal increases. When the phase shift reaches 180° the amplifier can become unstable and can even begin to oscillate. Oscillations render the amplifier circuit unsuitable for its intended purpose, since, during oscillations, the output of the amplifier bears no relationship to the AM signal applied to the amplifier's input. Therefore, in addition to the problem of being susceptible to noise at low output power levels, the envelope modulator circuit 108 in FIG. 3 is beset with stability concerns which limit the modulator's dynamic range.
Given the foregoing restrictions and limitations of the prior art, it would be desirable to have methods and apparatus for superimposing envelope information onto RF phase modulated signals in polar modulation transmitters that are efficient, stable, resistant to noise, and which are operable over wide dynamic ranges.